High frequency circuit analyser

ABSTRACT

An analyzer for measuring the response of an electronic device (DUT  206 ) to an RF input signal from a signal generator ( 240   a ) is described. An active load pull circuit ( 201 ) is connected to the DUT  206 , which receives an output signal from the DUT  206  and then feeds a modified signal back to the DUT  206 . The signal is modified by a signal processing circuit ( 237 ) in view of input signals x, y to control the magnitude gain and phase change effected by the feedback circuit ( 237 ). Thus, positive feedback loops are avoided and better control of the analyzer is permitted. A network analyzer, or other signal measuring device ( 242 ), logs the waveforms (from which s-parameters derived) observed at ports of the DUT  206 , thereby allowing the behaviour of the DUT  206  under various load conditions to be analyzed.

The present invention relates to analyzing the behaviour of a highfrequency device, in particular, a device for use in a high power (largesignal) high frequency amplifier, such as an amplifier for use in amobile telephone network or other telecommunications-relatedbase-station. The invention also relates to a method of improving theperformance of circuits including such a device.

When analyzing the behaviour of a high frequency electronic device it isoften desired to assess the behaviour of the device under the sort ofconditions that the device might be subjected to during normaloperation. For example, the impedance to which the device is attachedduring its normal/final operation may determine to a high degree theperformance, for example the efficiency and/or linearity, of the device.Such considerations are for example of particular relevance whendesigning high frequency large signal amplifier circuits for use in forexample a mobile telecommunications base station. It is thereforedesirous to be able to analyses the device when subjected to a virtualload/virtual impedance at the input and/or output of the device. Onemeans of applying such a virtual impedance is to apply an active loadpull, wherein a signal with a given magnitude and phase relative to aninput signal inputted into the device under test is injected into a port(for example the input or output) of the device under test.

A known load pull system is based on a feed-forward architecture, anexample of which being illustrated in FIG. 1 of the accompanyingdrawings. FIG. 1 shows a Device Under Test (DUT) 106 connected to a loadpull circuit 101 consisting of a power splitter 102, a phase shifter103, a variable attenuator 104, and an amplifier 105. Signals a_(SOURCE)received at the input side of the power splitter 102 are split into twosignals, one of which a′_(in) being fed via the phase shifter 103,variable attenuator 104, and amplifier 105 to one port of the DUT 106,resulting in the signal a_(out) (see the arrow a_(out) pointing fromright to left in FIG. 1 in accordance with standard convention), and theother a _(in) being fed directly to the other port of the DUT 106,producing the signal b_(out) (see the arrow b_(out) pointing from leftto right in FIG. 1 in accordance with standard convention). Thereflection coefficient Γ_(L), which determines the impedance seen by theDUT 106 is equal to the ratio of the travelling waves a_(OUT) andb_(OUT), such that Γ_(L)=a_(OUT)/b_(OUT). The reflection coefficientΓ_(L) is set by changing the magnitude and phase of the signal a_(OUT).

In the case of the feedforward load-pull circuit 101, the signal a′_(IN) entering the active load pull and the signal a_(OUT) leaving itare isolated when a quasi-unilateral device, for example an amplifier isused as DUT 106, keeping these two signals separated. If the input andoutput of the circuit are sufficiently isolated from each other then thecircuit can not form a feedback loop in which a signal circles withinthe loop, getting amplified with each pass through the loop, leading toan uncontrolled power build-up. Thus, by ensuring that the signala_(OUT) is isolated from the signal a′ _(IN) a stable operation of theactive load pull circuit 101 may be obtained. However, the system ofFIG. 1 suffers from a disadvantage in that the setting of the reflectioncoefficient Γ_(L) at the fundamental and the harmonic frequency is aniterative process. Since the signal a_(OUT) produced by the load pullcircuit is dependent on the signal a′ _(IN) and is independent ofb_(OUT), any change in the performance of the DUT 106, resulting in adifferent signal b_(out), for example power saturation, will introduce achange in the reflection coefficients Γ generated by the load pullcircuit 101. This makes the setting of the reflection coefficient Γdependent on the unknown behaviour of the DUT 106, which requirescontinual resetting of the settings of the phase shifter 103 andvariable attenuator 104 (by for example trial and error or by means of arandom search) in order to keep a constant reflection coefficient ateach power level. The need to adjust the settings of the phase shifter103 and variable attenuator 104 makes the feed-forward architectureunsuitable for automation.

It has been proposed to use feedback load pull circuits, but whilst theproposed circuits might not suffer from the above-mentioned disadvantageof the feed-forward architecture, problems associated with unstableoperation of such feedback load pull circuits have prevented suchproposals from being of any real use. Such instabilities result from theinput and output of the feedback circuit being unified, thus havingeffectively no isolation between them. It has been proposed to insertfilters into the feedback load-pull circuit, thereby separating theinput from the output of the circuit at most frequencies. However, atthe operational frequency of the filter there is still no effectiveisolation between the input and output of the load-pull, which mayresult in uncontrollable power build-up, which may then lead to signaloscillations rendering any measurements of the response of the DUT oflittle use.

The present invention thus seeks to provide an improved analyzer andmethod for analyzing the behaviour of an electronic device to a highfrequency input signal, and an improved method of designing andmanufacturing a high frequency device.

According to a first aspect of the invention there is provided ananalyzer for measuring at frequencies within a frequency range theresponse of an electronic device to a high frequency input signal, theanalyzer including:

an active load pull circuit connectable in use to a device to beanalyzed, the active load pull circuit including

a feedback circuit arranged to receive an output signal from the deviceto be analyzed, to modify the signal and to feed the modified signalback to the device to be analyzed, wherein

the feedback circuit is arranged to limit the magnitude gain of thefeedback circuit at all frequencies within the frequency range.

The output signal from the device to be analyzed may be received from aport of the device (for example an input port or an output port). Duringuse of the analyzer a high frequency signal may be applied to such aport of the device. Active load pull circuits may be used to greatadvantage when attempting to improve the design of a device or of acircuit in which the device is to be used. In order to enable suchimprovements to be made it is useful to be able to make measurementswhen the device is operating under conditions when the reflectioncoefficient is close to 1. The reflection coefficient is equal to theratio of the output signal at a certain frequency generated by theoutput from the. DUT (or the. wave passing from the DUT) to thereflected signal at the same frequency (or to the wave passing towardsthe DUT). Since the signal generated at the port of the DUT is a wave,often consisting of a number of frequencies the reflection coefficientwill generally be different at different frequencies. By means of thepresent invention it is possible to make measurements when thereflection coefficient is very close to 1, without the system becomingunstable. Without the ability to limit the magnitude gain of thefeedback circuit at all frequencies within the frequency range, theanalyzer (when connected to a DUT) might at certain frequencies withinthe frequency range form positive feedback loops (where the power gainof the circuit at that frequency might be greater than 1) potentiallyleading to system “lock-ups” and/or system instabilities. The functionof the feedback circuit of limiting the magnitude gain may thus beconsidered as effectively controlling the gain of the active load pullcircuit by altering (for example reducing) the bandwidth of the circuit(i.e. by limiting the magnitude gain to zero or close to zero atfrequencies outside the frequency range) and/or controlling its in-bandperformance (at frequencies within the frequency range).

In arriving at the present invention, it was recognized that theproblems relating to the instability of the active load pull feed backcircuits were due to the unexpectedly great variation in gain over arelatively narrow band of frequencies. Previously, it has been proposedto use band filters (such as YIG filters) in feedback load pullcircuits. The use of such filters mitigates to a limited extent thesystem instabilities that the present invention seeks to reduce oravoid. Such band filters typically had bandwidths of the order of 10%.Thus, frequencies across the bandwidth of the filter are able to createpositive feedback loops. Over this bandwidth, the amplifier employedwithin the active load pull circuit may have relatively large variationsin phase and magnitude, resulting in a largely varying power gain of thefeed back loop over the 10% bandwidth. Thus, such a circuit is generallyprone to generating positive feedback loops and consequently load pulloscillations. This problem becomes increasingly apparent when such acircuit is used to analyses the behaviour of a device when thereflection coefficient Γ_(L) at a given frequency within the bandwidthof the YIG filter is close to 1 (which is often necessary for theadequate characterisation of high power devices at 10 W or greater). Insuch cases, because the reflection coefficient Γ_(L)(=a_(OUT)/b_(OUT))≈1, the load-pull generates a signal a_(out) that isjust less than the signal b_(out) generated by the DUT at a firstfrequency at which the reflection coefficient is set at just below 1.Thus, the load-pull exhibits a signal gain close to 1 at the frequencyof primary interest. However, the gain of the feedback load-pull circuitat other frequencies close to the first frequency will also be close toone, due to the bandwidth of the filter. It is likely that at afrequency close to the first frequency the gain of the amplifier, orchange in its phase response, will be such that the reflectioncoefficient is larger than 1, which could resulting in oscillations atthis other frequency.

Thus whilst such use of band filters may mitigate the effects of systeminstabilities at low reflection coefficients (for example, significantlyless than 1), such a solution has limited application, because there areoften system instabilities when making measurements when the reflectioncoefficient is close to 1. It had not before the making of the presentinvention been appreciated that a problem with such systems laid in thelarge variation in gain over a relatively small bandwidth, making acontrol of the load-pull gain necessary inside as well as outside thebandwidth of operation (or the bandwidth of the band filter).

Furthermore, the device itself might be such that its behaviouroscillates. at the first frequency, even though the reflectioncoefficient is less than 1, thus leading to power build-ups andpotentially to damage to (and possibly destruction of) the device and/orthe analyzer. Such power build-ups at a frequency exactly equal to thefirst frequency cannot be avoided by the provision of a filter, becausesuch a filter would have to allow signals at the first frequency to pass(in order for the signals from the device at the frequency ofinterest—at the first frequency—to be generated and monitored). However,by limiting the in-band magnitude gain, for example by introducing acontrol of the magnitude and/or phase of the feedback circuit gain, theaforementioned problem can be avoided.

The present invention may also be of advantage in that it enables theprovision of an analyzer including an active load pull circuit, wherecomponents of the active load pull circuit need not have constant gain,even over a relatively narrow bandwidth, because the gain within thefrequency range of interest is effectively controlled. Thus thecomponents used in the active load pull circuit do not need to haveideal or close to ideal characteristics for the analyzer to performadequately. Indeed the components used may be substantially cheaper incost that might otherwise be required and thus costs may be reducedwithout greatly affecting overall performance.

Thus the invention provides an analyzer by which the analysis of highfrequency high power electronic devices may be made more reliable and/ormay be effected in a more cost-effective manner than hitherto possible.

The active load pull circuit may consist of the feedback circuit and noother components. However, the active load pull circuit mayalternatively include other components that do not contributesignificantly to the feedback effect of the feedback circuit.

The analyzer may be so arranged that the magnitude gain of the feedbackcircuit at one or more frequencies within the frequency range is able tobe adjusted.

The analyzer may be so arranged that the phase change effected by thefeedback circuit at one or more frequencies within the frequency rangeis able to be adjusted.

The feedback circuit may be arranged to restrict the phase changeeffected by the feedback circuit at all frequencies within the frequencyrange. For example, the feedback circuit may be arranged to restrict thephase change effected by the feedback circuit to minimise the risk ofpositive feedback in the feedback circuit and/or circuit oscillations.For example, phase changes that bring about such undesirable resultsgenerally relate to integer multiples of 180 degrees, for example 0degrees or 360 degrees.

According to a related aspect of the invention there is provided ananalyzer for measuring at frequencies within a frequency range theresponse of an electronic device to a high frequency input signal, theanalyzer including: an active load pull circuit connectable in use to adevice to be analyzed, the active load pull circuit including a feedbackcircuit arranged (i) to receive an output signal from the device to beanalyzed, (ii) to modify the signal and (iii) to feed the modifiedsignal back to the device to be analyzed, wherein the feedback circuitis arranged to control the magnitude and/or phase of the gain of thefeedback circuit at all frequencies within the frequency range. It willbe appreciated that any of the features described herein in relation toother aspects of the invention may be incorporated into this aspect ofthe invention.

The ability to control the magnitude and phase of the feedback atfrequencies in a certain bandwidth (the frequency range) by means of theanalyzer (and active load pull circuit) is a particularly advantageousfeature of the present invention. The feature enables an active loadpull circuit to be utilised in relation to signals at frequencies andpowers equal to those used in a real telecommunications system, wheresignals usually consist of closely spaced frequencies, which could notbe separated by filters (such as the YIG filters within the active loadpull circuits proposed in the prior art).

The analyzer is advantageously able to set a value of the impedance ateach of a plurality of frequencies and/or bandwidths. This may beachieved by means of the feedback circuit being able in use to apply apreset load to the device to be analyzed. The analyzer may be arrangedto be able to control the impedance at each of a plurality offrequencies/bandwidths. There may during use for example be a pluralityof load-pull circuits attached to the device. The analyzer may be ableto make measurements at frequencies outside the frequency range. Theanalyzer is preferably able to measure over a plurality of discreteranges of frequencies the response of an electronic device to a highfrequency input signal. The active load pull circuit, or the analyzer,may include a separate feedback circuit associated with each of theplurality of discrete ranges of frequencies. The or each feedbackcircuit is preferably arranged to limit the magnitude gain of thefeedback circuit at all frequencies within the frequency range withwhich the feedback circuit is associated. The frequency range, or atleast one of the discrete ranges of frequencies, may cover the frequencyof the input signal. The frequency range, or at least one of thediscrete ranges of frequencies, may for example be substantially centredon the frequency of the input signal applied in use to the device to beanalyzed. The discrete ranges of frequencies may each cover a frequencythat is a cardinal multiple of the input signal. The discrete ranges offrequencies may each be substantially centred on a frequency that is acardinal multiple of the input signal. It will be understood that thefrequencies corresponding to the cardinal multiples of the input signalwill include harmonics of the fundamental frequency of the input signal.

When the analyzer is arranged to measure over a plurality of discreteranges of frequencies, it may be considered that the frequency rangeincludes a plurality of discrete bands of frequencies. Such discretebands may be considered as sub-ranges of the frequency range.

The feedback circuit may also be arranged to modify signals outside thefrequency range. For example, the feedback circuit may be arranged tolimit the magnitude gain of the feedback circuit at certain frequenciesoutside the frequency range. Also, the feedback circuit may be arrangedto restrict the phase changes effected by the feedback circuit atcertain frequencies outside the frequency range. Such modificationsoutside of the frequency range need not necessarily be controllable.

The feedback circuit may be so arranged that it acts as a band filterhaving a bandwidth covering frequencies within the range. The feedbackcircuit may be so arranged that it acts as a band filter having abandwidth of greater than 10 MHz. The feedback circuit may, for example,include a high pass filter. The feedback circuit may, for example,include a low pass filter. The feedback circuit may, for example,include a band filter. Where there are more than one feedback circuitseach feedback circuit may include any the features described herein withreference to “the feedback circuit”.

The analyzer may include a high frequency band filter circuit arrangedto filter signals in or from the feedback circuit before they are fedback to the device, the band filter circuit having a bandwidth coveringfrequencies within the range. The feedback circuit may act as orcomprise a narrow band filter circuit. Said narrow band filter circuitmay for example simply be in the form of a narrow band filter, forexample forming a part of the feedback circuit. The gain of the feedbackcircuit may be such that between first and second frequencies differingby 1% there is a variation in the gain of the feedback circuit ofgreater than 5%. Said narrow band filter circuit may have a bandwidth ofless than 0.1% of the frequency on which the bandwidth is centred. Thecut-off frequencies of the bandwidth may both be between the first andsecond frequencies mentioned immediately above. Said narrow band filtercircuit may have a bandwidth such that the maximum variation in the gainof the circuit including the feedback circuit and said narrow bandfilter circuit in respect of frequencies within the bandwidth of saidnarrow band filter circuit is less than 20%.

The signal(s) applied to the device, in use, preferably include a signalhaving a fundamental frequency greater than 500 MHz. The analyzer isadvantageously suitable for analyzing high frequency devices whensubjected to high frequency signals, for example signals having afundamental frequency between 500 MHz and 50 GHz or more. Of course, theanalyzer may also be able to operate in respect of signals havingfrequencies outside this range.

The analyzer is advantageously suitable for analyzing high power deviceswhen subjected to high power signals, for example signals exceeding 1Watt and is especially advantageous at power levels exceeding 10 Watts.The device may be a high power transistor. The device may for example bea device suitable for use in a high power amplifying circuit in atelecommunications base station.

As mentioned above, the feedback circuit may act as a band filter forexample having a bandwidth of greater than 10 MHz (for example by meansof the feedback circuit comprising an appropriate filter such as a bandfilter). The feedback circuit need not comprise a band filter. In suchcases (where the feedback circuit performs a filtering function, butdoes not include a conventional band filter), it will be understood thatthe circuit may be considered as comprising a narrow band filtercircuit. The feedback circuit may be so arranged that it acts as a bandfilter having a bandwidth of greater than 20 MHz. Increasing theeffective bandwidth may increase the amount of information that can beascertained regarding a device to be analyzed by the analyzer. Having anarrow band filter, say having a bandwidth less than 10 Mz (atfrequencies of the order of GHz), may reduce problems associated withpositive feedback and circuit oscillations, but this is at the expenseof the information that can be ascertained regarding the device to beanalyzed with the analyzer. The filtering may, if over too narrow abandwidth, cut out frequencies of interest. Despite these disadvantages,there may in certain circumstances (for example in cases where it isdesired to reduce noise in the measurements) be advantage in having avery narrow bandwidth. The bandwidth of said narrow band filter circuitis preferably adjustable to bandwidths of less than 1 MHz. For example,if the frequency on which the bandwidth is centred is 1.8 GHz, thebandwidth may be of the order of 500 kHz and may even be of the order of200 KHz. Having such a narrow band filter may be of use in cases wherethe gain of the feedback circuit varies greatly with frequency. Forexample, the gain of the feedback circuit may vary by 10% over a 15 MHzbandwidth. The gain is therefore not constant over the relatively narrowbandwidth, but is well within acceptable limits. The analyzer may be soarranged that the maximum variation of the gain of the feedback circuit,during normal operating conditions, is less than 1% and more preferablyless than 0.1%. The bandwidth of the narrow band filter circuit may forexample be less than 0.05% of the frequency on which the bandwidth iscentred and may for example be of the order of 0.01% of the frequency onwhich the bandwidth is centred. The bandwidth of the band filter circuitis preferably variable, for example, between 0.05% and 10% of thefrequency on which the bandwidth is centred.

Advantageously, the frequency response of the feedback circuit is ableto be controlled and preferably able to be preselected. The narrow bandfilter circuit mentioned above could form any part of the active loadpull circuit, and does not necessarily form a part of the feedbackcircuit. The signal from the device to be analyzed may for example, passthrough-the narrow band filter circuit before or after being modified bythe feedback circuit.

The feedback circuit may include a heterodyne filter ring circuit. Theheterodyne filter ring circuit preferably includes a first mixer, asecond mixer, and a signal-modifying unit, preferably with variablebandwidth.

The heterodyne filter ring circuit is advantageously so arranged that inuse it receives an input at the first mixer together with a signalhaving a preselected frequency, and the output from the first mixer issent via the signal-modifying unit to the second mixer, where it iscombined with a signal having a frequency equal to the preselectedfrequency to produce the output signal of the heterodyne filter ringcircuit. The signal-modifying unit may be arranged to receive an inputsignal from the first mixer and to send an output signal to the secondmixer. The signal-modifying unit may comprise a signal processor.

The signal-modifying unit could include a digital signal processor thatis arranged to receive an analogue input signal from the first mixer viaan analogue-to-digital converter and to send an output signal to thesecond mixer via a digital-to-analogue converter.

Preferably, the signals having a preselected frequency received by thefirst and second mixers are produced by a single signal generator. Thesignal having a preselected frequency is advantageously produced by avariable signal oscillator. The signal-modifying unit may be in the formof, or act as, a band-pass filter, preferably with a variable bandwidth.The heterodyne filter ring circuit may thus be able effectively todown-convert an input signal to a lower frequency, to filter the signalat that lower frequency and to up-convert the filtered signal to ahigher frequency, the bandwidth of the heterodyne ring filter circuitbeing substantially equal to the bandwidth of the filtering at the lowerfrequency. It will be appreciated that the mixer(s) may be in the formof any suitable component that, when fed with two signals at differentfrequencies, outputs a signal including a component at a frequency equalto the difference between the frequencies of the input signals(preferably without any substantial non-linear behaviour at the outputfrequency under normal operating conditions).

The feedback circuit preferably includes a signal processor able in useto modify the signal from the device to be analyzed by a preselectableamount. The signal processor may be in the form of, or form a part of, asignal-modifying unit. For example, the signal processor may be in theform of a signal-modifying unit of the heterodyne filter circuitmentioned above. The signal-modifying unit may be an analoguesignal-modifying unit. The signal-modifying unit may be a digitalsignal-modifying unit. In the case, where the signal-modifying unit (orsignal processor) is arranged to receive and/or to output a digitalsignal, there is advantageously provided a converter (either or both ananalogue-to-digital converter and a digital-to-analogue converter, asappropriate).

Preferably, the signal processor is adjustable, so that the modificationof the signal from the device is able to be altered. The signalprocessor is advantageously programmable, for example so that themodification of the signal from the device is able to be pre-programmed.

The signal processor may be in the form of a variable amplitudemodifying circuit. The signal processor may be in the form of a variablephase modifying circuit able in use to modify the phase of a signal fromthe device to be analyzed by a preselectable amount. Preferably, thesignal processor is able to modify both the phase and magnitude ofsignals.

In an embodiment described below of the invention, the signal-modifyingunit includes a digital signal processor that is arranged to receive ananalogue input signal from the first mixer via an analogue-to-digitalconverter and to send an output signal to the second mixer via a digitalto analogue converter. In such a case it is preferred that the digitalsignal processor is in the form of a computer or is operated under thecontrol of a computer. In that embodiment the analyzer includes a narrowband filter, and the analogue-to-digital converter is an 8 bit samplerand samples the incoming analogue signal at a rate of at least fourtimes the frequency on which the bandwidth of the narrow band filter iscentred. The digital signal processor, and possibly theanalogue-to-digital converter and/or the digital to analogue converter,is/are advantageously so arranged that, in use, the analogue signaloutputted by the digital to analogue converter is filtered to excludecomponents of the signal received at the analogue-to-digital converteroutside a given bandwidth, thereby performing the function of a bandfilter.

The signal processor, especially when in the form of a digital signalprocessor, may be arranged to compensate for non-ideal behaviour ofcomponents of the analyzer and, in particular, of the active load pullcircuit. For example, when a heterodyne filter ring circuit is provided,the (optionally digital) signal processor may be arranged to compensatefor non-linear behaviour of the mixers. Such a use of the digital signalprocessor may provide a less expensive solution to problems associatedwith mixers, or other components, having unacceptable non-linearbehaviour, rather than simply replacing the components with betterquality, and more expensive components. The (optionally digital) signalprocessor may also be arranged to compensate for problems associatedwith signal leakages, for example of the signal generator(s) used toprovide the signals having a preselected frequency. Signal leakagescould alternatively or additionally be reduced by providing a furtherfilter arranged to block such signal leakage.

The signal processor is advantageously arranged to provide both in-bandand out-of-band signal modification. For example, the signal processormay be arranged to filter out signals outside a given band offrequencies and to control the magnitude-and/or-phase of signals withinsaid band so as to reduce the likelihood of positive feedback (with theaim of avoiding signal oscillations within the load-pull circuit).

The feedback circuit advantageously includes a variable amplitudemodifying circuit able in use to modify the amplitude of a signal fromthe device to be analyzed by a preselectable amount. The variableamplitude modifying circuit may for example include a variableamplifier. The variable amplitude modifying circuit may include avariable amplifier and a fixed amplifier. The amplitude modifyingcircuit may be able in use to modify the amplitude of a signal from thedevice to be analyzed by an amount dependent on the frequency of thesignal, or components thereof.

The feedback circuit advantageously includes a variable phase modifyingcircuit able in use to modify the phase of a signal from the device tobe analyzed by a preselectable amount. A signal modulator may, forexample, form at least a part of the phase modifying circuit. Thevariable phase modifying circuit may be able in use to modify the phaseof a signal from the device to be analyzed by an amount dependent on thefrequency of the signal, or components thereof.

The (optionally digital) signal processor (for example of the heterodynering filter circuit) described above may form at least a part of avariable amplitude modifying circuit and/or form at least a part of avariable phase modifying circuit. Advantageously, the (optionallydigital) signal processor is able, in use to act as a filter circuit, anamplitude modifying circuit and a phase modifying circuit. The functionof the amplitude and phase modifying circuits may be provided by thedigital signal processor by means of it processing the IQ values of thesignal.

The signal processor may be arranged to process respective signalsrepresentative of the I and Q values of a signal. The feedback circuitmay include an IQ-modulator and an IQ demodulator. For example, thesignal processor mentioned above may receive signals from an IQdemodulator and send signals to an IQ modulator. The signal processorcould, for example, by processing the IQ signals perform the function ofboth an amplitude and a phase modifying circuit. The IQ modulator and/ordemodulator may be arranged to be under the control of a computer. TheIQ demodulator may be arranged to receive an input signal having at agiven frequency an amplitude and a phase and to produce two outputsignals, one relating to the I-value and the other relating to theQ-value, the I and Q values being representative of the point on arectangular plot corresponding to the amplitude and phase. The IQmodulator may be arranged to receive two input signals, one relating toan I-value and the other relating to a Q-value, the I and Q values beingrepresentative of the point on a rectangular plot corresponding to anamplitude and a phase and to produce an output signal at the givenfrequency, having the amplitude and phase corresponding to the I and Qvalues. The IQ modulator and demodulator may modulate and demodulatesignals having a plurality of frequency components. The IQ modulator andIQ demodulator may each comprise one or more signal mixers.

The feedback circuit may comprise a component, comprising for example amixer or an IQ demodulator, that converts a high frequency input signalinto a signal that is able to be processed by the signal processor. Insuch a case, the feedback circuit may comprise a component, comprisingfor example a mixer or an IQ modulator, that converts the signalprocessed by the signal processor back into a high frequency outputsignal.

The feedback circuit may for example be arranged to downconvert highfrequency input signals to a lower frequency, modify the signals, andthen upconvert the modified signals to high frequency output signals.Thus, the feedback circuit may operate at frequencies significantlydifferent from the frequency of the signal(s) fed back to the device tobe analyzed. The active load pull circuit may therefore, in use, producean output signal at a frequency significantly different from thefrequency at which the feedback circuit is controlled. Oscillationsbetween the device to be analyzed and the active load pull circuit canbe significantly reduced, or eliminated, if there are no frequencies atwhich the device to be analyzed and the active load-pull circuit caninfluence each other, such as frequencies the same as or similar to thefrequency of the signal which is fed back to the device to be analyzedby the active load-pull circuit, (typically, a high frequency signal,for example of the order of 1.8 GHz). However, any internal oscillationsin the active load pull circuit are likely only to occur at thedownconverted frequencies at which the signals are modified (typically,a relatively low frequency signal, for example of the order of 20 kHz).Consequently, any variations of the impedance of the device to beanalyzed can be observed only at 1.8 GHz, which are highly unlikely togive rise to any oscillations at 20 kHz. As a result, the separation ofthe frequency at which the active load pull produces the feedback signalfrom the frequency at which the signal is modified provides stabilityand greatly reduces the risk of internal circuit oscillations andpositive feedback at all frequencies of interest. Such an advantage mayhave independent application and as such the present invention furtherprovides an analyzer for measuring at frequencies within a frequencyrange the response of an electronic device to a high frequency inputsignal, the analyzer including a load pull circuit connectable in use toa device to be analyzed, the load pull circuit including a signalmodifying circuit arranged (i) to receive a signal to be modified, (ii)to downconvert the signal received to a low frequency signal, to modifythe low frequency signal, to upconvert the modified low frequency signalto a modified high frequency signal and (iii) to feed the modifiedsignal to the device to be analyzed. Such an arrangement may reduce therisk of undesirable positive feedback or signal oscillations. Thisaspect of the present invention may of course include features of otheraspects of the invention described herein. For example, the load pullcircuit may be in the form of a feedback circuit, so that the signalreceived by the signal modifying circuit is one that is outputted fromthe device to be analyzed. The downconversion of signals may result inthe signal being converted into I and Q signals as described above.

The analyzer preferably includes a signal generator arranged to send aninput signal to the device to be analyzed. The signal generator ispreferably able to produce high frequency signals of, for example, atleast 1GHz. The signal generator is preferably able to produce highfrequency signals of, for example, up to 50 GHz. The frequency of the oreach component of the signal produced by the signal generator isadvantageously able to be preselected.

The analyzer preferably includes a signal-measuring device for measuringloads arising in response to the signals applied to the device to beanalyzed. The signal-measuring device may, for example, be in the formof a vector network analyzer or a microwave transition analyzer.

The analyzer advantageously includes a computer for controlling and/orautomating processes. The computer is preferably programmed to be ablein use to set substantially all of the parameters able to be preselectedby the electronic components of the analyzer. Preferably a singlecomputer is provided to perform the running of the analyzer and thelogging of data resulting from measurements made during the running ofthe analyzer.

Above, reference is made to a heterodyne filter ring circuit includingfirst and second mixers, and a digital signal processor that may act asa band filter circuit. Such a filter ring circuit may have applicationsother than in-relation to performing the function of modifying signalsin accordance with the present invention. The present invention thusprovides a filter circuit comprising a first mixer and asignal-modifying unit, wherein the filter circuit is so arranged that,in use, the circuit receives a first input signal at the first mixertogether with a second signal produced by a part of the filter circuit,the first mixer combines the first and second signals to produce a thirdsignal having a component at a difference frequency, the differencefrequency being equal to the difference between the frequencies of thefirst and second signals, and the third signal is modified by thesignal-modifying unit substantially to remove any components of thethird signal at frequencies outside a band of frequencies that includesthe difference frequency. Advantageously, the resulting modified signaloutputted by the signal-modifying unit is then received at a secondmixer, where it is combined with a fourth signal produced by a part ofthe filter circuit to produce a filtered output signal retainingcharacteristics of the input signal. Advantageously, the filter circuitis arranged to receive and output high frequency signals, in which casethe first, second and fourth signals may be high frequency signals.Preferably the frequency of the second signal is substantially equal tothe frequency of the fourth signal.

Preferably, the magnitude difference between the second signal and thefourth signal is substantially constant, and may for example be zero (orat least negligible). Preferably, the phase of the second signal issubstantially equal to the phase of the fourth signal. The second andfourth signals are conveniently produced by the same part of the filtercircuit, which may for example be in the form of a variable signalgenerator. The third signal advantageously retains information fromwhich the phase and magnitude of the first signal can be ascertained.The characteristics retained in the output signal of the circuit may forexample relate to the phase and/or the magnitude of the input signal.Advantageously, the circuit may be used to receive a signal having acomponent at a given frequency, having a given phase and magnitude, andto output a signal substantially consisting of that component and havingthe same phase and magnitude. Advantageously, the filter circuit may beused as a very narrow band high frequency filter circuit where thepass-band has a width less than 0.1% (and more preferably less than0.05%) of the frequency on which the band is centred, which frequency ispreferably greater than 500 MHz. Advantageously, the filter bandwidthmay be variable to adjust to the receiving first signal.

The present invention also provides an active load pull circuit suitablefor use as the active load pull circuit of the analyzer of the presentinvention as described herein. The active load pull circuit may forexample include a feedback circuit arranged to receive an output signalfrom the device to be analyzed, to modify the signal and to feed themodified signal back to the device to be analyzed, wherein the feedbackcircuit is arranged to limit the magnitude gain of the feedback circuitat all frequencies within the frequency range. The active load pullcircuit may also include any of the features of the active load pullcircuit of the analyzer of the present invention as described herein.

The present invention also provides a method of measuring the responseof an electronic device to a high frequency input signal, the methodincluding the steps of:

providing an electronic device to be analyzed,

applying a high frequency signal to the device, and

modifying an output signal from the device and then feeding the modifiedsignal back to the device, thereby forming a feedback loop, and

measuring, at a plurality of frequencies within a frequency range, theresponse of the device to the signal applied to the device,

wherein the magnitude gain of the feedback loop is limited atfrequencies within the frequency range.

The method may include a step of preselecting the way in which theoutput signal from the device is modified. For example, the method mayinclude a step of programming a control unit, microprocessor or thelike. The modification of the output signal from the device may beperformed such that the signal is modified in a different way atdifferent frequencies. The modification of the output signal may involvemodifying the phase and/or the magnitude of the signal. The phase changeeffected by the feedback loop may be restricted at certain frequencieswithin the frequency range. The method may include a step ofpreselecting the magnitude of gain applied to the output signal from thedevice. The method may include a step of preselecting a phase changeapplied to the output signal from the device.

The measuring of the response of the device to the signal applied to thedevice may be performed at a multiplicity of frequencies across a singlerange of frequencies. The measuring of the response of the device to thesignal applied to the device may be performed at a plurality ofdifferent frequencies across a plurality of ranges of frequencies. Themeasuring of the response of the device to the signal applied to thedevice may be performed in respect of a plurality of frequencies withinany of a plurality of discrete frequency ranges. The ranges offrequencies may each correspond to fundamental frequency of the signalapplied to the device and its harmonics. The or each discrete frequencyrange may be centred on a frequency that is a cardinal multiple of thefrequency of the fundamental frequency of the signal applied to thedevice. The load pull method may, if the number is 2 or more, thus be inthe form of a harmonic load pull method. The magnitude gain of thefeedback loop may be limited at frequencies within a plurality ofdiscrete frequency ranges.

The step of modifying the output signal from the device may includefiltering out signals having frequencies outside a band of frequenciescovering frequencies within the frequency range. The step of modifyingthe output signal from the device may include filtering out all signalshaving frequencies within a plurality of discrete sub-ranges offrequencies. The sub-ranges of frequencies may cover frequencies betweenthe upper and lower frequencies defining the frequency range.

The method is advantageously able to be performed such that the gain ofthe feedback loop at frequencies within the frequency range is limitedto be less than 1. The method may be performed in such a way that thegain within the frequency range varies from below 0.5 to above 0.8. Themethod may be performed in such a way that the gain at the frequenciesof interest is between 0.5 and 1, possibly between 0.8 and 1.

The fundamental frequency of the signal applied to the device ispreferably over 1 GHz. The fundamental frequency may between 500 MHz and50 GHz.

The method is advantageously repeated and performed in respect of amultiplicity of different modifications of the output signal from thedevice. The method may for example be repeated and performed in respectof a multiplicity of different input signals applied to the device. Thedifferent input signals applied to the device may be at differentfrequencies and/or under different load conditions. The method may forexample be performed to simulate different loads on the device. Themethod may be performed to simulate different impedances applied to thedevice. A load or impedance may be simulated by means of the differencebetween the signal applied to the device and the corresponding signalresponse from the device. Of course, at least one port of the devicewill be subjected to an active load pull.

The modifications made to the signal on each separate performance of themethod may effectively consist of a systematic trace of signals throughthe IQ plane. The modifications made may include a multiplicity ofdifferent modifications of the I value at each of a multiplicity ofdifferent modifications of the Q value. For example, the modificationsmade may include at least ten different modifications of the I value ateach of at least ten different modifications of the Q value.

The input signal is preferably a high power input signal of, forexample, at least 1 Watt and more preferably greater than 10 Watts. Themethod is advantageously repeated and performed in respect of amultiplicity of different input signals applied to the device. Thedifferent input signals may for example be a multiplicity of separateinput signals at different powers. The input signals may for example bea multiplicity of separate input signals at different frequencies. Foreach input signal, the method is preferably repeated and performed inrespect of a multiplicity of different modifications of the outputsignal from the device being analyzed. For example, for each inputsignal, the multiplicity of different modifications made to the signalfed back to the device may effectively consist of a systematic trace ofsignals through the IQ plane (for example, as described above).

The step of modifying the output signal may be performed by a circuitincluding a heterodyne filter ring circuit. The heterodyne filter ringcircuit may be in the form of the heterodyne filter ring circuitaccording to any of the aspects of the analyzer of the present inventionas described herein. For example, the heterodyne filter ring circuit mayinclude a first mixer, a second mixer, and a signal-modifying unit. Themethod may be such that the first mixer receives an input signal at afirst frequency together with a signal having a second frequency, thesecond frequency being preselected to be close to the first frequency,the output from the first mixer being sent to the signal-modifying unit,the signal-modifying unit extracting a signal having a frequency equalto the difference in the frequency of the first and second frequencies,and outputting a signal derived from the extracted signal to the secondmixer, where it is combined with a signal having a frequency equal tothe second frequency to produce the output signal of the heterodynefilter ring circuit. The signal-modifying unit may be in the form of thesignal-modifying unit according to any of the aspects of the analyzer ofthe present invention as described herein. It is especially preferred,for example, that the signal-modifying unit includes a signal processorthat advantageously processes signals representative of IQ values or thelike. The second frequency may be preselected to be substantially equalto the first frequency, so that the signal extracted by thesignal-modifying unit is a DC signal. In such a case, the method may beso performed that information regarding the magnitude of the signalreceived at the first mixer can be ascertained from the magnitude of theDC signal and the phase of the signal received at the first mixerrelative to the phase of the output signal from the second mixer can becontrolled by changing the relative phase of the signal having thefourth frequency with respect to the phase of the signal applied to thedevice. Other features of the heterodyne ring circuit described hereinwith reference to the analyzer of the present invention may, whereappropriate, be incorporated into the aspects of the method of theinvention as described above. For example, the signal-modifying unit ofthe filter circuit may be in the form of a band-pass filter and/or mayinclude a signal processor, for example a digital signal processor.

A signal processor is advantageously used in the method, for example, inthe step of modifying the output signal from the device. The signalprocessor may perform a band filtering function. The signal processormay perform one or more of the following steps: modifying the frequencyof the signal, modifying the magnitude of the signal, modifying thephase of the signal, modifying the I-value of the signal and the Q-valueof the signal. The signal processor may also compensate for non-idealbehaviour of one or more other components used in the method.

The present invention also provides according to a related aspect of theinvention (which may be incorporated into the step of modifying anoutput signal as described above) a method of filtering and/or modifyingan input signal having a first frequency, which is preferably greaterthan 500 MHz, the method including the steps of combining the inputsignal with a signal at a second frequency to produce a third signalhaving a component at a difference frequency, the difference frequencybeing equal to the difference between the first and second frequencies,and modifying the third signal substantially to remove any components ofthe third signal at frequencies outside a band of frequencies thatincludes the difference frequency, wherein the band has a width, whichcan be less than 0.1% (and preferably less than 0.05%) of the firstfrequency. Preferably the method also includes a step of combining thethird signal as modified with a signal having a fourth frequency, thefourth frequency preferably being substantially equal to the secondfrequency. Such a method may be incorporated into any aspect of themethod of the present invention.

The method preferably includes one or more calibration steps. One suchcalibration step may be performed to calibrate the loads generated bythe feedback loop in relation to the amount of modification made to thesignal fed back to the output of the device. During such a calibrationstep the load generated by the feedback loop, in relation to a givenmodification made to the signal fed back to the output of the device, isadvantageously measured and recorded. A multiplicity of differentmodifications are advantageously made during the calibration,measurements of the loads generated being recorded in respect of eachsuch signal modification. During subsequent performance of the method,it is then advantageously possible, during the step of modifying thesignal, for an appropriate modification to be made in order to produce apredetermined load at the output of the device. A user or a controllingcomputer may thus be able to select a desired load (to be applied by theactive load pull), which is then applied by means of the automaticselection of the appropriate signal modification in view of thecalibration data. For example, the method may be such that thecalibration data relates to loads produced by a multiplicity of signalmodifications representable by a matrix of points in the IQ plane, acomputer is instructed to apply a desired load by means of the activeload pull, and the computer ascertains the I and Q values that wouldproduce the desired load (by interpolation or extrapolation ifnecessary).

The invention also provides a calibration method comprising repeatingthe following steps for a multiplicity of different loads:

applying a high frequency signal at the input of the feedback loop orfeedback circuit, and

modifying the applied high frequency signal and feeding the modifiedsignal back to the input to synthesise a load,

measuring, at a plurality of frequencies within a frequency range, themodified signal at the input,

calculating the load represented by the feedback loop or feedbackcircuit in response to the particular modification made to the appliedsignal, and storing electronically the results of the measurementsagainst the modifications to the signal. The calibration method may beperformed so that predetermined loads are applied at the output of thedevice by selecting an appropriate modification during the step ofmodifying the signal in accordance with the electronically storedmeasurements.

The method is advantageously performed at least partly under the controlof a computer.

The method may be performed with an analyzer according to the presentinvention as described herein. The analyzer or active load pull circuitof the present invention as described herein is preferably so arrangedas to be able to perform any or all aspects of the method of the presentinvention as described herein.

The present invention further provides a method of improving the designof a high frequency high power device or a circuit including a highfrequency high power device, the method including the steps of analyzingthe behaviour of the device either by using the analyzer according tothe present invention or by performing the method according to thepresent invention, and then modifying the design of the device ormodifying the circuit including the device in consideration of theresults of the analyzing of the behaviour of the device. The presentinvention yet further provides a method of manufacturing a highfrequency high power device or a circuit including a high frequency highpower device, the method including the steps of improving the design ofa similar existing device or of an existing circuit including such adevice by performing the method described immediately above and thenmanufacturing the device or the circuit including the device inaccordance with the improved design.

Reference is made herein to limiting the magnitude gain of the feedbackcircuit at all frequencies within the frequency range. It will beappreciated that the limiting of the gain need only be actively appliedat frequencies at which the gain is likely to give rise to positivefeedback or oscillations. Thus, where a signal processor or othercomponent or device is used actively to limit the magnitude gain, such acomponent or device need only actively limit the magnitude gain atcertain frequencies within the frequency range, the limiting of themagnitude gain of the feedback circuit at other frequencies within thefrequency range being provided as a natural consequence of thearrangement of the feedback circuit in relation to the other devices(such as the DUT, for example) and/or components to which it isattached.

It will be understood that any features of the above-described aspectsof the invention may be incorporated into other aspects of the presentinvention. For example, features described with reference to theanalyzer of the invention may be incorporated with suitable changes intoaspects of the method of the present invention.

Embodiments of the invention will now be described, by way of exampleonly, with reference to the accompanying schematic drawings of which,

FIG. 1 shows a circuit diagram of a prior art active load pull circuit;

FIG. 2 a shows a schematic circuit diagram of an active load pullcircuit according to a first embodiment;

FIG. 2 b shows a schematic circuit diagram of an active load pullcircuit according to a second embodiment;

FIG. 3 shows a narrow band filter suitable for use in the circuit shownschematically in FIG. 2 a;

FIG. 4 shows a modulator suitable for use in the circuit shownschematically in FIG. 2 a;

FIG. 5 is a graph of the gain of the circuit of FIG. 1, without a narrowband filter, against frequency;

FIG. 6 shows a combined narrow band filter and modulator for use in athird embodiment of the invention;

FIG. 7 shows a modulator for use in a fourth embodiment of theinvention;

FIG. 8 shows points in an IQ plane at which measurements are made duringcalibration;

FIGS. 9 a and 9 b show results of s-parameter measurements made at thepoints shown in FIG. 8;

FIG. 10 shows a further embodiment of the invention featuring a signalprocessing unit;

FIG. 11 shows the signal processing unit of FIG. 10 in further detail;and

FIGS. 12 a and 12 b show Smith charts showing results of the measurementof s-parameters using the embodiment shown in FIGS. 10 and 11.

FIG. 2 a shows a schematic circuit diagram in accordance with a firstembodiment of the invention, the diagram showing a feedback active loadpull circuit 1 connected to a DUT 6 (Device Under Test). The DUT may forexample be a high power transistor, such as a “LDMOS ” (laterallydiffused metal-oxide silicon) device. The circuit 1 consists of anamplifier, a signal circulator 7, and means for manipulating signalsinside a frequency band and outside a frequency, the means beingrepresented schematically by a band filter 8, and a signal modifier 9.

The signal modifier 9 allows the phase and magnitude of signals to bealtered and allows the active load pull applied by the circuit 1 to theDUT 6 to be controlled. In use signals b_(OUT) enter the feedbackcircuit 1 from the DUT 6, are then fed from the signal circulator 7 tothe filter 8, then to the signal modifier 9, and then to the amplifier5, the amplified signals then passing back via the signal circulator 7to the DUT 6 as signals a_(OUT). The signal a_(OUT) produced by the loadpull circuit 1 is dependent on the signal b_(OUT) produced by the DUT 6,and therefore changes in the signal b_(OUT) cause corresponding changesin the signal a_(OUT) (provided that the characteristics of thecomponents of the load pull circuit do not change significantly; forexample the saturation of the amplifier remains substantially constant).Thus, in use, the reflection coefficient Γ_(L) (the ratioa_(OUT)/b_(OUT)) is effectively locked once it has been set, and doesnot vary significantly with changes in the loading or biasing of the DUT6.

The filtering and manipulation of the signals performed by the feedbackcircuit 1 (represented schematically by filter 8 and modifier 9) ensuresa stable operation of the feedback load pull system. The measurementsmade with the load pull system are all concerned with signals withinknown frequency ranges, for example, frequencies at or around thefundamental or a given harmonic frequency. Frequencies outside theseranges, and which are therefore of no interest, may be filtered out (thefiltering being represented by band filter 8, but it will of course beappreciated that other means could perform such a filtering step).Without filtering, signals at other frequencies might cause systeminstabilities. For example, because the isolation between the signalb_(OUT) entering the load pull circuit 1 and the signal a_(OUT) leavingthe circuit is provided only by the reflection coefficient of the DUT 6,there is a possibility of oscillations occurring at a frequency withinthe bandwidth of the load pull components as soon as the gain of theloop is larger than one (i.e. leading to the formation of a positivefeedback loop at one or more frequencies). Whilst, such positivefeedback loops can be avoided at the harmonic frequency by setting thecharacteristics of the variable components of the active load pullcircuit, feedback loops could still occur at other frequencies becauseof the great variance in gain of the circuit with a modest change infrequency. Signals within the frequency ranges of interest are notfiltered out by the filter 8, but are prevented from causing positivefeedback by means of signal-modifying unit 9 performing in-band signalmanipulation (for example, by attenuating the signal at givenfrequencies within the range).

A graph illustrating (in respect of a circuit excluding the band filter8 and modifier 9, but otherwise being identical to that of FIG. 2 a) howgain varies with frequency can be seen in FIG. 5. At a frequency of 1.8GHz a variation in frequency of only 15 MHz can cause about 0.8 dB(almost 20%) change in gain. Thus if the load pull circuit operates in acondition close to positive feedback at the frequency of interest, therewould, without the band filter 8 and signal-modifying unit 9, existpositive feedback loops at other frequencies, which coupled togetherwith inherent “noise” in the system would quickly render the systemunstable. At a frequency of 1.8 GHz a variation in frequency of 200 KHz(i.e. a 0.01% bandwidth) still produces a 0.05 dB (about 1%) change ingain. Whilst the effective gain of the amplifier 5 need not be constantacross the bandwidth of the feedback circuit 1, if the filter 8 were tobe used alone to avoid a positive feedback situation, the bandwidthwould have to be so narrow that measurements of the response of the DUT6 would have to be very significantly limited.

The frequency on which the bandwidth of the filter 8 is centred is ableto be pre-selected thus enabling the load pull circuit 1 to be used at avariety of different frequencies. The ability to so tune the filter 8also allows the amplifier to have a bandwidth significantly wider thanthe bandwidth of the filter.

FIG. 2 b shows a schematic circuit diagram in accordance with a secondembodiment of the invention, the circuit being in accordance with theschematic diagram of the first embodiment (FIG. 2 a). The circuitincludes a feedback active load pull circuit 1 connected to a DUT 6. Thecircuit 1 consists of an amplifier 5, a signal circulator 7, andsignal-modifying means 9 for manipulating signals inside a frequencyband and outside a frequency band. The signal-modifying means 9 includesan IQ demodulator 36, which receives signals from the signal circulator7.The signals from the IQ demodulator 36 are received by a digitalsignal-processing unit 37 (which may be in the form of a PC, orspecially configured digital signal processor). The processed signalsfrom the digital signal-processing unit 37 are received by an IQmodulator 38.In use, the signal b_(out), generated by the DUT 6, is fedvia the signal circulator 7, to the IQ demodulator 36.The IQ demodulatorgenerates the signal I′ and Q′, which represent the magnitude-and phaseof the signal b_(out) in rectangular co-ordinates. The I′-valuerepresents the x-value and the Q′-value represents the y-value of thesignal b_(out) on a rectangular xy-plot. The IQ demodulatordownconverts, by means of a combination with a signal from a localoscillator source 39, the I′ and Q′ signals to a frequency, which issufficiently low to be able to be processed by the digital signalprocessing unit 37, which digitises and then modifies the I′ and Q′signals. The I′ and Q′ signals are modified to ensure that no positivefeedback loop is caused at a frequency within a given range. Themodified I and Q values are then fed to the IQ modulator 38, whichgenerates a signal with a magnitude and phase represented by the I and Qsignals. The signal generated by the IQ modulator 38 has the samefrequency as the signal b_(out) (the IQ modulator 38 upconverts themagnitude and phase information contained within the I and Q signals tothe higher frequency, by means of a combination with a signal from alocal oscillator source 39). The signal outputted by the IQ modulator 38is then fed through the amplifier 5 into the DUT output. The digitalsignal-processing unit 37 effectively filters the signals. The change inthe input and output IQ values of the input and output signals I′ and Iand Q′ and Q generates a difference between the b_(out) and a_(out)waves, thereby facilitating the control of the reflection coefficientΓ=a_(out)/b_(out).

Converting the magnitude and phase information contained within thesignal b_(out) to a lower frequency, and producing I′ and Q′ signals,has the advantage that this information can be digitised and thenmanipulated by the re-programmable digital circuitry 37.The digitalcircuitry 37 can vary the effective bandwidth of the filtering of the I′and Q′ signals performed by the feedback circuit 1. The bandwidth may bevaried in dependence on the frequency content of the signal b_(out).Also the offset between I′ and I as well as Q′ and Q can be variedreadily, thus controlling the frequency response of the load pull withinthe bandwidth of the filtering. As a result, the frequency response ofthe load-pull circuit may be controlled both outside and inside thebandwidth of the signal a_(out) and b_(out).

The band filter 8 shown schematically in FIG. 2 a may alternatively bein the form of a heterodyne filter circuit, shown in more detail in FIG.3. The signals entering the band filter 8 are represented by arrow 13 aand the signals leaving the band filter 8 are represented by arrow 13 b.The filter 8 comprises a tuneable local oscillator (LO) 11 that in useprovides signals to two mixers 10 a and 10 b, interposed between whichthere is a conventional band-pass filter 12.The conventional bandfilter, which is in the form of a surface acoustic wave filter (SAWfilter), has a range of between 169.9 MHz and 170.1 MHz (i.e. abandwidth of 200 kHz). In use the local oscillator is set to produce twoidentical sine-wave signals (in phase with each other), represented byarrows 16, at a frequency close to the first harmonic frequencygenerated by DUT 6 such that the two frequencies are separated by 170MHz. For example, if the fundamental frequency of the harmonic signalgenerated by the DUT is at 1.8 GHz the Local oscillator is set tooscillate at 1.63 GHz. The first mixer 10 a sums the two signals andoutputs a signal including a component having a frequency equal to thedifference in frequencies of the two signals inputted at the mixer 10 a,which in this case is 170 Mhz. Thus the incoming signal (arrow 13 a)from the DUT 6 is effectively down-converted to a lower frequencysignal, which is represented by arrow 14. The down-converted signalsthen pass through the conventional band-pass filter 12 and then (arrow15) into the second mixer 10 b, into which the other of the two signals(arrow 16) from the local oscillator 11 is also passed. The mixer 10 bthus outputs a signal (arrow 13 b) essentially consisting of a componenthaving a frequency equal to the signal originally passed into the filterall signals having frequencies close to that frequency but more than 200KHz apart having been filtered out. Thus the filter circuit 8effectively acts as a very narrow band filter centred on a frequency of1.8 GHz and having a bandwidth of 200 KHz. It will be appreciated thatthe pass filter 8 is tuneable by means of the Local Oscillator 11.

Of course it will be appreciated that if the load pull measurements areto be made over a frequency range having a 10% bandwidth, a filtercircuit such as that shown in

FIG. 3 may not be appropriate as such a circuit might filter out signalshaving frequencies outside the bandwidth of the filter circuit 8, butstill being of interest. In such cases, the bandwidth of the filtercircuit can be widened and the signal modulation capability of thesignal-modifying unit 9 used to modify signals within the frequencyband, thereby effectively actively flattening the frequency response ofthe load pull circuit over the bandwidth.

The signal modifier 9 shown schematically in FIG. 2 a may be in the formof an IQ modulator simply controlled by a computer as shown in FIG. 4.In this embodiment, the signal modifier 9 comprises a programmedcomputer 17 and an IQ modulator 18. The IQ modulator 18 receives signals19 from the band filter 8 (not shown in FIG. 4). The phase and magnitudeof the signals entering the IQ modulator are changed in accordance withI and Q values set by the computer 17 the resultinq output signal 20being passed to the amplifier 5 (not shown in FIG. 4). The computer 17can sweep through many I and Q values in sequence to provide resultsspanning an I-Q plane. Signals from the computer 17 are converted intoDC signals by a digital to analogue converter (not shown) beforeentering the DC controllable I and Q inputs of the IQ modulator 18.Again, the computer 17 is able to perform in-band modification of thesignals and the filter circuit 8 (not shown in FIG. 4) performs out-bandfiltering of the signals.

Before, analyzing the characteristics of a DUT, it is first necessary tocalibrate the active load pull circuit 1. During calibration, thecomputer 17 causes the IQ modulator 18 to step through I and Q valuesand, at each point in the IQ plane, the load generated by the load pullcircuit 1 is measured at measurement reference plane A (see FIG. 2 a)with a VNA (vector network analyzer), such as for example aHewlett-Packard HP_(—)8753 VNA. The VNA also produces an input signal,which is fed into the active load pull circuit 1.(The VNA effectivelyreplaces the DUT 6 as shown in FIG. 2 a.) Measurements are made inrespect of IQ points across a matrix of 121 ×121 points as shown in FIG.8. According to this present embodiment, the points at whichmeasurements are made reach only a magnitude of up to 0.4 of the inputsignal. However, it will be appreciated that the magnitude of the signaloutputted by the amplifier 5 may have a magnitude in excess of 0.4depending on the gain of the amplifier 5.The calibration process takesabout 10 minutes to perform. From the loads measured at the points inthe I-Q plane, two contour plots are generated in the (s-parameter) s₂₁load plane, one plot having contours representing constant I values (theQ-values changing as along the length of each contour), the other plothaving contours representing constant Q values (the I-values changing asalong the length of each contour). It is thus possible to generate anydesired load/impedance, within the boundaries of the contour plotsproduced, by setting I and Q values appropriately. Any loads/impedances,which were not generated by the load pull during the calibrationprocess, may be reproduced by interpolating between the points availableon the contour plots. Example, contour plots produced by such acalibration step are shown in FIGS. 9 a and 9 b.

During subsequent analysis of a DUT, measurements are made at referenceplane B, by means of a Microwave Transition Analyzer (MTA) connected toa directional coupler interposed between the active load pull circuit 1and the DUT 6. The introduction of a coupler alters the previouslycalibrated network, where measurements were made in respect of referenceplane A. Thus, as a further step in the calibration procedure, a 3 pointcalibration is conducted to enable the loads at reference plane B (theoutput port of the DUT 6) to be mapped to corresponding values atreference plane A to enable the loads set by the active load pullcircuit 1 to relate to loads at reference plane B. The load pull circuit1 is attached via a directional coupler (not shown) to a pre-calibratedmeasurement system in the form of a microwave transition analyzer (MTA)(not shown) which is able to measure signals derived from a_(out) andb_(out), Then 3 “known” loads are applied by the load pull circuit, forexample, by setting the I and Q values to [0.2, 0], [0, 0] and [−0.2,0]. From the difference between the loads measured by the measurementsystem, at reference plane B, and the known loads previously measured atplane A by the VNA during the previous calibration procedure, thes-parameters of the network between planes A and B can be extracted.Thus, the loads set by the load pull circuit can be set at valuesrelating to the loads at plane B (at the DUT output port) as opposed toplane A.

The calibration and control of the active load pull circuit is automatedby means of a suitably programmed computer. The software used in thepresent embodiment provides the ability to set voltage ranges (in the IQplane) and the amount of points at which the calibration process isperformed. After the calibration step has been performed the programmedcomputer is able to apply a load equivalent to any reflectioncoefficient within the Smith chart, once provided with details of adesired magnitude and phase by the user. The programmed computer canalso be instructed to perform a reference plane calibration, if a shiftof the reference plane is necessary (enabling the user to specifydesired reflection coefficients in respect of the new reference planeB).

In use the fully calibrated analyzer including the active load pullcircuit 1, MTA and directional coupler is connected to a DUT 6, thecoupler being interposed between the DUT and the load pull circuit. Asignal generator then produces an input signal a at a pre-selectedfrequency which is applied to the input port of DUT 6. The load pullcircuit 1 then applies different pre-selected load pulls andmeasurements of the response of the DUT 6 are made with the MTA. Thewhole process is governed by a single suitably programmed computer.

The programmed computer also facilitates the automation of more complexload pull measurements. For example, measurements may be made at amultiplicity of different input signals a_(IN) (received by the DUT 6).The computer sets an input signal and then causes the load applied bythe active load pull to be set at a pre-selected value. The computerthen receives input data relating to the measurement made at theparticular input signal and the active load pull applied. Thereafter theload applied by the active load pull is changed from one value to thenext in a sequence that spans the region of interest in the IQ plane andat each load a further measurement is made. For example, measurementsmay be made over a range of magnitudes from 0 to 1 at a resolution of0.2 and at a phase resolution of 5 degrees (a total of 360 points in theIQ plane arranged as five concentric load circles). After measurementshave been made in respect of a sufficient number of points in the IQplane, the input signal is then changed and the process repeated. Sincethe feedback load pull tracks the output signal generated by the DUT nocorrections are necessary in order to keep the load constant for varyingpower levels. Measurements may for example be made in respect of amultiplicity (20, for example) of different power levels of inputsignals spanning a range of 0 to 30 dBm in steps of 0.1 dBm (i.e. 1 mWto 1W). Changing from one set of parameters to another and performingthe load measurements can be completed with the aid of the computerwithin a relatively short period of time (in a fraction of a second).Thus a sufficient number (for example 7,200) of load pull measurementsto characterise a DUT can, with the present embodiment, be conductedwithin a period of time of the order of minutes. Such results can thenbe utilised to improve the design of power amplifiers, for example,power amplifiers for use in telecommunication base stations.

Measurements carried out to determine the accuracy of the load pullsystem have given a maximum magnitude error of 0.006 and a maximum phaseerror of 0.5°.

Thus the first embodiment provides an active load pull measurementsystem for characterising an electronic device (to enable theimprovement of the device and circuits in which the device is to beused), wherein a lower quality amplifier may be used without prejudicingthe overall performance of the system. Costs may thereby be reduced.

According to a third embodiment of the invention, the function of theband filter 8 and signal modifier 9 (shown in FIG. 2 a illustrating thefirst embodiment) are provided by means of a digital signal processingcircuit 21, shown in FIG. 6. The circuit 21 comprises a tuneable localoscillator (LO) 22 that in use provides signals to two mixers 23 a and23 b, interposed between which there are an eight-bitanalogue-to-digital converter (ADC) 24, a digital signal processor 25,and an eight-bit digital to analogue converter 26. (The bit-resolutionof the ADC and the DAC can of course be increased if greater accuracy isrequired.) The signals entering the circuit represented by arrow 27 andthe signals leaving the circuit 21 are represented by arrow 28. In usethe local oscillator is set (by under the control of the DSP 25) toproduce two identical sine-wave signals (in phase with each other),represented by arrows 29, at a frequency very close to the firstharmonic frequency generated by DUT 6 in a manner similar to that of thenarrow band filter 8 described with reference to FIG. 3. The first mixer23 a sums the two signals and outputs a signal including a componenthaving a frequency equal to the difference in frequencies of the twosignals inputted at the mixer 23 a, which in this case is 1 Mhz. Thusthe incoming signal (arrow 27) from the DUT 6 is effectivelydown-converted to a lower frequency signal, which is represented byarrow 30. The down-converted signals then passes to the ADC, whereuponthe signal is converted to a digital signal. The sampling rate is set tobe at least four times the frequency of the incoming signal. In thepresent embodiment the sampling rate of the ADC is 40 MHz, whilst thefrequency of the signal being sampled is 1 MHz. The DSP then processesthe signal and outputs a digital signal that is then converted into ananalogue signal by the DAC 26. The resulting output signal 31 is thenpassed to the second mixer 23 b and is recombined with the other of thetwo signals (arrow 29) from the local oscillator 22 at the second mixer23 b. The output signal 31 from the DAC 26 contains substantially nosignal components at frequencies other than the frequency of interest.

It will be appreciated that the DSP can be arranged to process thesignals received to improve the performance of the load pull circuit 1.The DSP both performs a filtering function and an in-band signalmanipulation function. The DSP is thereby able to reduce the chance ofpositive feedback loops existing in the load pull circuit. Also,components of the circuit 1 may have non-ideal behaviour and the DSPcould be programmed to compensate for such non-idealities. For example,the mixers may have behaviour that whilst non-linear is readilycharacterised. Once characterised (during a suitable calibrationprocedure) compensations may be made by the DSP in respect of thatnon-linear behaviour.

According to a fourth embodiment of the invention shown in FIG. 7, thesignal modifier 9 of the first embodiment is replaced with a simplesignal modulator circuit 32 comprising a variable phase shifter 33 and avariable amplifier 34, both of which are controlled by a suitablyarranged computer.

According to a fifth embodiment, which utilises the apparatus andcircuits provided in accordance with the third embodiment, the LO iscaused to oscillate at the same frequency as the frequency of interest.The output of the first mixer 23 a therefore comprises a DC componentrepresentative of the magnitude of the component of the input signal(arrow 27) at the frequency of interest. Control of the relativemagnitude and phase of the load applied by the active load pull iscontrollable by modifying the phase of the LO relative the phase of thesignal inputted into the DUT 6 and generated by the single generator(not shown) and by modifying the magnitude of the DC signal sent to thesecond mixer 23 b. It will be appreciated that the same technique couldbe used in relation to the first and fourth embodiments.

A sixth embodiment is illustrated in FIGS. 10 and 11. The circuit shownschematically in FIG. 10 is similar in concept to that of the secondembodiment and comprises a DUT 206 connected at a first port to a firstsignal source 240 a (in this case, in the form of an Agilent ESG RFsignal generator available from Agilent Technologies Inc, a UScorporation). A second port of the DUT 206 is connected to an activeload pull circuit 201. The load pull circuit extracts a small fractionof the signal travelling from the second port of the DUT 206 by means ofa first signal coupler 241, the extracted signal being fed to an IQdemodulator 236, which effectively downconverts the input signal into Iand Q signals. The signals from the IQ demodulator 236 are received byan analogue signal-processing circuit 237, which is described below infurther detail. The analogue signal-processing circuit 237 transformsthe I and Q signals received from the IQ demodulator into new signals I′and Q′, by means of a preset transformation defined in part by inputsignals x and y. The transformed signals are received by a second signalgenerator 240 b (also an Agilent ESG RF signal generator), whicheffectively upconverts the I′ and Q′ signals into a modified feedbacksignal (with a magnitude and phase represented by the I′ and Q′signals), which is fed back to the second port of the DUT via the firstcoupler. Effectively, the signal generator 240 b is used here as an IQmodulator, by means of the in-built IQ modulator of the signal generator240 b. Thus, the circuit 201 acts as an active load pull feedbackcircuit with the feedback loop being controllable by means ofcontrolling the inputs x and y supplied to the signal-processing circuit237. The circuit of the sixth embodiment may be considered as effectinga transfer function which controls the relationship between the signalgenerated by the DUT 206 and the signal generated by the active loadpull circuit 201, the transfer function being able to be controlled bythe programmable DC sources providing the signals x and y. Measurementsare made by means of a oscilloscope 242, which is supplied with signalsrepresentative of the transmitted and reflected signals at the firstport of the DUT by means of second and third couplers 243 a and 243 b(provided in series and between the first signal generator 240 a and theDUT 206) and signals representative of the transmitted and reflectedsignals at the second port of the DUT by means of fourth and fifthcouplers 244 a and 244 b (provided in series and between the firstcoupler 241 signal generator 240 a and the DUT 206).

FIG. 11 shows the signal-processing circuit 237 of the sixth embodimentin further detail. DC signals, x and y, generated by two separatecontrollable signal generators 245 x and 245 y and input signals I andQ, represented by box 246, from the IQ demodulator 236 (not shown inFIG. 11) are fed into four signal multipliers 247 a, 247 b, 247 c and247 d to produce signals representative of xI, yQ, xQ and yI,respectively. The signal representative of yQ is fed via an inverter toa first signal summation device 248 a, which also receives the signalrepresentative of xI. Thus, the output of the summation device isrepresentative of xI-yQ. The signals representative of xQ and yI are toa second signal summation device 248 b, which outputs a signalrepresentative of xQ+yI. The output of the first and second summationdevices are fed as transformed signals I′ and Q′ (represented by box249) to the signal generator (not shown in FIG. 11), which effectivelyupconverts and modulates the I′ and Q′ signals.

Thus the transformation function, F(x,y) effected by the analoguesignal-processing circuit 237 can be expressed asI′=IX−QY; andQ′=IY+QX, such that: $\begin{matrix}{{F_{x,y}\left( {I,Q} \right)} = {I^{\prime} + {j\quad Q^{\prime}}}} \\{= {\left( {{x \cdot I} - {y \cdot Q}} \right) + {j\left( {{x \cdot Q} + {y \cdot I}} \right)}}} \\{{= {\left( {x + {j\quad y}} \right)*\left( {I + {j\quad Q}} \right)}},{and}}\end{matrix}$  (I′+jQ′)/(I+jQ)=x+jy=Z,where Z is a constant complex number.

Choosing Z as a constant results in the relationship between the signalgenerated by the device under test (DUT 206) and the signal generated bythe load pull circuit 237 also remaining constant. This means that anychange in the signal is reproduced by the load pull circuit, thussimulating a constant impedance. This is of particular use for modulatedsignals, which exhibit varying signal levels over the modulationbandwidth (in much the same way as occurs within telecommunicationsystems such as GSM or UMTS).

FIGS. 12 a and 12 b show some of the results obtained with themeasurement configuration shown in FIGS. 10 and 11. For thesemeasurements a signal with three tones was 15 used with a centrefrequency at 1.8 GHz and a tone separation of 20 kHz. (It will beappreciated that larger tone separations can be used.) The first plotshown in FIG. 12 a was produced by setting the magnitude of the loadsignal to be half that of the input signal, then sweeping its phase in90° steps. Since the load presented is broad band, i.e. it is constant,the three points lie on top of each other for each phase setting. In thesecond plot shown in FIG. 12 b the impedance is swept in a circle aroundthe smith chart with a measurement taken every degree, which is achievedby sweeping values of X and Y. In the plot illustrated by FIG. 12 b, twocircles of points may be discerned, the first lower circle 250representing the carrier and the second upper circle 251 representingthe plot points for the upper and lower sidebands, which are so close toone another in FIG. 12 b as to be indistinguishable from one another.

In summary, the analyzer measures the response of the DUT 206 to an RFinput signal (which may be multi-tone) from the signal generator 240 a.The active load pull circuit 201, connected to the DUT 206, receives anoutput signal from the DUT 206 and then feeds a modified signal back tothe DUT 206. The signal is modified by the signal rocessing circuit 237,in view of input.signals x, y, to control the magnitude gain and phasechange effected by the feedback circuit 237. The signal measuring device(oscilloscope 242) measures the waveforms (from which s-parameters canbe derived) observed at ports of the DUT 206, thereby allowing thebehaviour of the DUT 206 under various load conditions to be analyzed.Positive feedback loops are able to be avoided and better analysis ofthe behaviour of the DUT is made possible by means of the better controlafforded by the load pull circuit 237. It will be readily apparent tothe skilled person that various modifications may be made to theabove-described embodiment without departing from the spirit of theinvention. For example, if desired, the DSP 25 described above withreference to the second embodiment may also be arranged, under thecontrol of a suitably programmed computer (not shown in FIG. 3), toprocess the incoming signal 30 to modify its magnitude and phase,thereby removing the requirement for a separate modulator.

The bandwidth of the very narrow band filter circuit formed by means ofthe circuit shown in FIG. 3 may be narrower than 200 KHz, which at aninput signal of 2 GHz represents a bandwidth of 0.01%. It is envisagedthat it would be possible to reduce the bandwidth to as low a figure asis required. The minimum bandwidth that it is proposed would beimplemented in accordance with the above-described embodiments is thelower of 200 kHz or 0.01% of the frequency on which the narrow band iscentred.

It will be understood that, in relation to the embodiment illustrated byFIG. 3, the filtering characteristics of the band filter circuit 8 aredetermined by the choice of conventional filter 12 and the relativefrequency difference between the input signal and the frequency of thesignal produced by the LO 11. Once a conventional band pass filter 12has been chosen (having the desired bandwidth), the frequency on whichthe band filter 8 is centred can be set by pre-selecting, an appropriatefrequency signal to be produced by the LO 11. For example, if theconventional filter has a 100 KHz bandwidth centred at 1 MHz and thedesire is to centre the bandwidth of the very narrow band filter at 1.8GHz, the LO 11 would be set to oscillate at 1.799 GHz or 1.801 GHz.

Also, features of one embodiment may, where appropriate, be readilyincorporated in another embodiment. For example, the filter 8 and thesignal modifier 9 of the schematic circuit of first embodiment may beformed by the DSP circuit 21 of the third embodiment and the modulator32 of the fourth embodiment, respectively.

Other improvements to the above-described embodiments may be made. Forexample, with reference to the circuit shown in FIG. 3, depending on thequality of the mixers used, there may be some unwanted signal leakageback into the DUT from the active load pull-circuit. Such leakage mayfor example originate from the Local Oscillator. If so, and if the LocalOscillator is set to oscillate at a frequency different from that of thefrequency of interest such signal leakages can be mitigated by means ofa further filtering circuit, for example a high pass filter or low passfilter with a cut-off frequency between the LO signal frequency and theoutput frequency of the load pull circuit. LO leakage may alternativelybe compensated for, in the second embodiment, by means of the DSP.

Since the signal generated by the output port of the DUT is a wave,often consisting of a number of frequencies, the output port effectivelysees a separate, generally different, reflection coefficient at eachfrequency. The load pull systems shown in the Figures manipulate areflection coefficient at a single frequency or a single continuousfrequency band. The value of the impedance, i.e. load, at each frequencyand/or bandwidth may be controlled by attaching for eachfrequency/bandwidth a separate load pull circuit to the device. Such amulti-tone analyzer may be provided, for example, consisting of aplurality of active load pull circuits having band filters centred ondifferent frequencies (for example, at different harmonic frequencies).Each active load pull circuit could for example be in the form of thecircuit 1 of FIG. 2 b, the circuit being connected to the DUT via asignal junction or a signal splitter. Alternatively, one such load pullcircuit (as shown in FIG. 2 b) could include a signal splitter arrangedto receive signals from the signal circulator 7, the split signals beingfed via respective separate portions of the circuit, each portionincluding a filter, a modulator and an amplifier.

One modification of the sixth embodiment that has been envisaged relatesto the case where a control of the signals over a wider bandwidth isdesired. In that case, rather than using DC sources to generate the xand y values, arbitrary waveform generators (AWGs) could be used,thereby permitting the control of the x and y values over a widerbandwidth.

In the embodiments described above, the load pull circuit has been shownas being connected to the output only of the DUT. There could thereforebe provided an analyzer, for example, consisting of at least two activeload pull circuits, one load pull circuit being provided at the input tothe DUT and the other being provided at the output.

1. An analyzer for measuring at frequencies within a frequency range theresponse of an electronic device to a high frequency input signal, theanalyzer including: an active load pull circuit connectable in use to adevice to be analyzed, the active load pull circuit including a feedbackcircuit arranged (i) to receive an output signal from the device to beanalyzed, (ii) to modify the signal and (iii) to feed the modifiedsignal back to the device to be analyzed, wherein the feedback circuitis arranged to limit the magnitude gain of the feedback circuit at allfrequencies within the frequency range.
 2. An analyzer according toclaim 1, wherein the analyzer is so arranged that the magnitude gain ofthe feedback circuit at one or more frequencies within the frequencyrange is able to be adjusted.
 3. An analyzer according to claim 1,wherein the analyzer is so arranged that the phase change effected bythe feedback circuit at one or more frequencies within the frequencyrange is able to be adjusted.
 4. An analyzer according to claim 1,wherein the feedback circuit is arranged to restrict the phase changeeffected by the feedback circuit at all frequencies within the frequencyrange.
 5. An analyzer according to claim 1, wherein the feedback circuitis so arranged that it acts as a band filter having a bandwidth coveringfrequencies within the range.
 6. An analyzer according to claim 1,wherein the analyzer includes a high frequency band filter circuitarranged to filter signals in or from the feedback circuit before theyare fed back to the device, the band filter circuit having a bandwidthcovering frequencies within the range.
 7. An analyzer according to claim5, wherein the feedback circuit is so arranged that it acts as a bandfilter having a bandwidth of greater than 10 MHz.
 8. An analyzeraccording to any preceding claim 1, wherein the feedback circuitincludes a heterodyne filter ring circuit.
 9. An analyzer according toclaim 8, wherein the heterodyne filter ring circuit includes a firstmixer, a second mixer, and a signal-modifying unit, the heterodynefilter ring circuit being so arranged that in use it receives an inputat the first mixer together with a signal having a preselectedfrequency, and the output from the first mixer is sent via thesignal-modifying unit to the second mixer, where it is combined with asignal having a frequency equal to the preselected frequency to producethe output signal of the heterodyne filter ring circuit.
 10. An analyzeraccording to claim 1, wherein the feedback circuit includes a signalprocessor able in use to modify the signal from the device to beanalyzed by a preselectable amount.
 11. An analyzer according to claim10, wherein the signal processor is arranged to process respectivesignals representative of the I and Q values of a signal.
 12. Ananalyzer according to claim 1, wherein the analyzer includes a signalgenerator arranged to send an input signal to the device to be analyzed.13. An analyzer according to claim 1, wherein the analyzer includes asignal measuring device for measuring loads arising in response to thesignals applied to the device to be analyzed.
 14. An active load pullcircuit for use in an analyzer for measuring at frequencies within afrequency range the response of an electronic device to a high frequencyinput signal, the active load pull circuit being connectable in use to adevice to be analyzed and including a feedback circuit arranged toreceive an output signal from the device to be analyzed, to modify thesignal and to feed the modified signal back to the device to beanalyzed, wherein the feedback circuit is arranged to limit themagnitude gain of the feedback circuit at all frequencies within thefrequency range.
 15. An active load pull circuit according to claim 14,wherein the active load pull circuit is so arranged that at least one of(a) the magnitude gain of, and the (b) phase change effected by, thefeedback circuit at one or more frequencies within the frequency rangeis able to be adiusted.
 16. A method of measuring the response of anelectronic device to a high frequency input signal, the method includingthe steps of: providing an electronic device to be analyzed, applying ahigh frequency signal to the device, and modifying an output signal fromthe device and then feeding the modified signal back to the device,thereby forming a feedback loop, and measuring, at a plurality offrequencies within a frequency range, the response of the device to thesignal applied to the device, wherein the magnitude gain of the feedbackloop is limited at frequencies within the frequency range.
 17. A methodaccording to claim 16, wherein the phase change effected by the feedbackloop is restricted at frequencies within the frequency range.
 18. Amethod according to claim 16, wherein the method includes a step ofpreselecting the way in which the output signal from the device ismodified.
 19. A method according to claim 18, wherein the methodincludes a step of preselecting a magnitude gain applied to the outputsignal from the device.
 20. A method according to claim 18, wherein themethod includes a step of preselecting a phase change applied to theoutput signal from the device.
 21. A method according to claim 16,wherein the step of modifying the output signal from the device includesfiltering out signals having frequencies outside a band of frequenciescovering frequencies within the frequency range.
 22. A method accordingto claim 16, wherein the fundamental frequency of the signal applied tothe device is over 1 GHz.
 23. A method according to claim 16, whereinthe method is repeated and performed in respect of a multiplicity ofdifferent modifications of the output signal from the device.
 24. Amethod according to claim 16, wherein the method is repeated andperformed in respect of a multiplicity of different input signalsapplied to the device.
 25. A method of calibrating an analyzer accordingto claim 16, wherein the calibration method comprising repeating thefollowing steps for a multiplicity of different loads: applying a highfrequency signal at the input of the feedback loop or feedback circuit,and modifying the applied high frequency signal and feeding the modifiedsignal back to the input to synthesise a load, measuring, at a pluralityof frequencies within a frequency range, the modified signal at theinput, calculating the load represented by the feedback loop or feedbackcircuit in response to the particular modification made to the appliedsignal, and storing electronically the results of the measurementsagainst the modifications to the signal.
 26. A method according to claim16, wherein the method includes performing a calibration, theperformance of the calibration comprising repeating the following stepsfor a multiplicity of different loads: applying a high frequency signalat the input of the feedback loop or feedback circuit, and modifying theapplied high frequency signal and feeding the modified signal back tothe input to synthesise a load, measuring, at a plurality of frequencieswithin a frequency range, the modified signal at the input, calculatingthe load represented by the feedback loop or feedback circuit inresponse to the particular modification made to the applied signal, andstoring electronically the results of the measurements against themodifications to the signal, so that predetermined loads may be appliedat the output of the device by selecting an appropriate modificationduring the step of modifying the signal in accordance with theelectronically stored measurements.
 27. A method according to claim 16,wherein the method is performed with ananalyser, the analyzer including:an active load pull circuit connectable in use to a device to beanalvsed, the active load pull circuit including a feedback circuitarranged (i) to receive an output signal from the device to be analyzed,(ii) to modify the signal and (iii) to feed the modified signal back tothe device to be analyzed, wherein the feedback circuit is arranged tolimit the magnitude gain of the feedback circuit at all frequencieswithin the frequency range.
 28. A method of improving the design of ahigh frequency high power device or a circuit including a high frequencyhigh power device, the method including the steps of analyzing thebehaviour of the device by using the analyzer of claim 1, and thenmodifying the design of the device or modifying the circuit includingthe device in consideration of the results of the analyzing of thebehaviour of the device.
 29. A method of manufacturing a high frequencyhigh power device or a circuit including a high frequency high powerdevice, the method including the steps of improving the design of asimilar existing device or of an existing circuit including such adevice by performing the method of claim 28 and then manufacturing thedevice or the circuit including the device in accordance with theimproved design.
 30. An analyzer for measuring at frequencies within afrequency range the response of an electronic device to a high frequencyinput signal, the analyzer including: an active load pull circuitconnectable in use to a device to be analyzed, the active load pullcircuit including a feedback circuit arranged (i) to receive an outputsignal from the device to be analyzed, (ii) to downconvert the signalreceived to a low frequency signal, to modify the low frequency signal,to upconvert the modified low frequency signal to a modified highfrequency signal and (iii) to feed the modified signal back to thedevice to be analyzed, wherein the feedback circuit is arranged to limitthe magnitude gain of the feedback circuit at all frequencies within thefrequency range.
 31. An analyzer for measuring the response of anelectronic device to a high frequency input signal, the analyzerincluding: an active load pull circuit connectable in use to a device tobe analyzed, the active load pull circuit including a feedback circuitarranged (i) to receive an output signal from the device to be analyzed,(ii) to modify the signal, the modification including limiting themagnitude gain of the feedback circuit at frequencies outside a band offrequencies, (iii) to feed the modified signal back to the device to beanalyzed, the modified signal fed back comprising a component having afrequency within said band, and (iv) the feedback circuit is alsoarranged to limit the magnitude gain of the feedback circuit atfrequencies inside the band of frequencies.
 32. A method of improvingthe design of a high frequency high power device or a circuit includinga high frequency high power device, the method including the steps ofanalyzing the behaviour of the device by performing the method of claim16, and then modifying the design of the device or modifying the circuitincluding the device in consideration of the results of the analyzing ofthe behaviour of the device.
 33. A method of manufacturing a highfrequency high power device or a circuit including a high frequency highpower device, the method including the steps of improving the design ofa similar existing device or of an existing circuit including such adevice by performing the method of claim 32 and then manufacturing thedevice or the circuit including the device in accordance with theimproved design.